SK201B is a 13VDC signal trigger output which is active
whenever the amplifier is powered up. R218 and DZ207 / C223
provide a reference voltage which is buffered by TR200. TR201
and R217 act as a current limit and prevent damage due to a short
circuit on the output of SK201B. The maximum current is
approximately 65mA.
TR203 and TR202 are a complementary Darlington pair which
turn on mains relay RLY200 when activated by a signal from the
microprocessor.
TR204 and its associated components are to detect whenever AC
mains is present at the IEC socket. This is to notify the
microprocessor if the user has unplugged the mains cord, so that it
can take the necessary action (muting all the outputs and switching
off the mains relay). The reservoir capacitors should last at least 4
mains cycles which gives the microprocessor plenty of time for a
controlled shutdown.
TR204 forms a monostable circuit. Each cycle of AC turns on
TR204 via R211. TR204 then ‘shunts’ C229 ensuring that it is
kept at a low potential. If more than one mains cycle is missing,
then R219 charges up C229 sufficiently to trigger Schmitt inverter
IC202E thus passing on a logic signal to the microprocessor. The
use of a Schmitt inverter for IC202 is to ensure that the micro
receives ‘clean’ logic levels - the hysteresis voltage (about 0.5V) is
sufficient to prevent circuit noise from producing a string of
‘ghost’ signals when analogue levels are near the threshold point.
TH200 is a positive tempco thermistor placed adjacent to the
heatsink on which the output transistors are mounted. When the
temperature of the thermistor exceeds 90 degrees Celsius the
thermistor goes to a high impedance and so the input to IC202F
goes low. This triggers a HIGH output to the micro indicating
thermal overload.
The VI protection signals from the left and right channels pass into
IC202A and IC202B respectively, to be ‘cleaned up’ via the
Schmitt trigger. They are then NOR’d using TR205 which sends a
HIGH signal to the micro in the event of either channel suffering a
short circuit or current overload. Exactly the same approach is
used for the DC fault lines using IC202C and IC202D.
L882 Circuit Sheet 3
This is the main audio power amplifier circuit. The amplifier is a
class B design, which uses SAP ‘audio’ transistors in a
symmetrical current feedback configuration. Input and feedback
paths are DC coupled and there is an active integrating servo to
remove DC offsets from the output.
The basic principle of operation is follows:
The input signal is amplified by a factor of 2 in IC300A. This
drives a 44 impedance to ground causing the supply pin currents
to change with the signal level. These changing supply pin currents
are then ‘reflected’ by a pair of complementary Wilson mirrors
and passed on to a series of buffer transistors before being
connected to the load. The ‘feedback current’ flows back from the
output terminal via R331 and R332 and attempts to provide the
current necessary to allow IC300A to swing its output without
drawing excessive current from its supply pins, thus making the
change in supply current very small indeed. This is why the term
‘current feedback’ is used - it is the current flowing in the
feedback resistors that sets the overall gain of the amplifier.
IC300B acts as an inverting integrator and its purpose is to remove
DC from the loudspeaker output. Any positive DC offset will
cause the output of IC300B to go negative, thus increasing the
current in its negative supply pin and pulling the output voltage
back towards zero. R330 and C317 set the time constant of this
integrator (0.47 seconds) so that audio frequency components are
ignored and only DC and subsonic frequencies are removed.
The input to the amplifier is limited to ±5.4V via back-to-back
zener diodes DZ302 and DZ303. This is to prevent the user from
grossly overdriving the input to the amplifier and possibly causing
damage. The diodes appear before series resistor R324 so that their
variable capacitance does not introduce high frequency harmonic
distortion.
R324, R327 and C316 act as an input filter - this is a first order
low pass filter with a corner frequency of around 340kHz to
prevent RF signals from being injected into the front end of the
amplifier. The corner frequency was chosen such that the phase
shift introduced is less than 5 at 20kHz (considered by the AES
to be the minimum perceptible relative amount by the human ear).
The input impedance of the amplifier is 23kW at DC, falling to
around 14kW at 20kHz.
Operational amplifier IC300A is acting as a non-inverting gain of
2, driving the input signal into a 44W impedance to ground via
R322 and R337. Its output voltage will be an accurate
amplification of its input voltage (i.e. the signal on pin 1 should
look identical to that on pin 3 but at twice the amplitude). The op-
amp is used in a slightly unusual configuration here, in that its
power supply pins are used as a (current) output, and its output pin
is used as a (current) feedback.
Transistors TR311 and TR303 supply the ±15V rails to the op-
amp, and act as cascades to pass its supply pin currents through to
the current mirrors, which sit at a potential too high for the op-amp
to be connected directly.
TR300, TR301 and TR321 form a PNP Wilson current mirror,
which reflects the current sunk by the positive supply pin of
IC300. Likewise TR314, TR315 and TR320 form an NPN Wilson
current mirror, which reflects the current sourced by the negative
supply pin of IC300.
R315 thru R318 provide emitter degeneration of approximately
300mV for the current mirrors (as they pass about 3mA DC in
quiescent conditions), to ensure accurate operation independent of
the small variations between the transistors in the current mirrors.
They also ensure that the current passing down the next stage is
reasonably constant as the internal temperature of the amplifier
changes, swamping out small thermal variations in the V
BE
of the
mirror transistors.
R319 and R320 slightly decouple the rails to the current mirrors
from the main power rails of the amplifier, to allow the bootstrap
circuit to operate. The bootstrap consists of C302 and C306 with
metal film power resistors R352 and R353. The bootstrap is
provided to allow the power supply rails of the current mirrors to
go up and down slightly with the output signal into the
loudspeaker. This enables the driver stage to fully saturate the
output transistors and thus give the greatest power output and best
thermal efficiency for any given power rail voltage. The voltage on
the ‘inside’ end of R319 and R320 will vary by about 12 volts
peak to peak at full output power, rising above the main power
rails during signal peaks.
C307 and C308 with R333 and R335 provide the compensation
necessary to ensure stability when the loop is closed. They are
Miller capacitors which dramatically reduce the transimpedance
(i.e. current to voltage gain) of the current mirrors at high
frequencies. The present value of 47pF provides for a unity gain
open loop bandwidth of around 75MHz, whilst ensuring a closed
loop gain margin of around 6dB (note that gain margin in a current
feedback design is not dependent on system bandwidth to a first
order approximation). R333 and R335 provide a ‘zero’ in the open
loop frequency response which is tailored to give the best time
domain performance (i.e. to make high frequency square waves
look square with minimal ringing or overshoot).
DZ304 and C311 provide a fixed 4.7V bias voltage to allow the
following stages to operate correctly. C311 is there to ensure that
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